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1 LT1578/LT1578-2.5 1.5a, 200khz step-down switching regulator 3.3v buck converter efficiency vs load current n 1.5a switch current n high efficiencylow loss 0.2 w switch n constant 200khz switching frequency n 4v to 15v input voltagerange n minimum output: 1.21v n low supply current: 1.9ma n low shutdown current: 20 m a n easily synchronizable up to 400khz n cycle-by-cycle current limit n reduced emi generation n low thermal resistance so-8 package n uses small low value inductors the lt ? 1578 is a 200khz monolithic buck mode switching regulator. a 1.5a switch is included on the die along with all the necessary oscillator, control and logic circuitry. the topology is current mode for fast transient response and good loop stability. the LT1578 is a modified version of the lt1507 that has been optimized for noise sensitive appli- cations. it will operate over a 4v to 15v input range. in addition, the reference voltage has been lowered to al- low the device to produce output voltages down to 1.2v. quiescent current has been reduced by a factor of two. switch on resistance has been reduced by 30%. switch tran- sition times have been slowed to reduce emi generation. the oscillator frequency has been reduced to 200khz to maintain high efficiency over a wide output current range. the pinout has been changed to improve pc layout by al- lowing the high current, high frequency switching circuitry to be easily isolated from low current, noise sensitive con- trol circuitry. the new so-8 package includes a fused ground lead that significantly reduces the thermal resistance of the device to extend the ambient operating temperature range. standard surface mount external parts can be used including the inductor and capacitors. n portable computers n battery-powered systems n battery chargers n distributed power systems , ltc and lt are registered trademarks of linear technology corporation. load current (a) 0 efficiency (%) 90 85 80 75 70 65 60 55 50 0.25 0.50 0.75 1.00 1578 ta02 1.25 1.50 v out = 3.3v v in = 5v l = 25 h descriptio u features applicatio s u boost LT1578 v in shdn output** 3.3v, 1.25a * ripple current rating 3 i out /2 ** increase l1 to 30 m h for load currents above 0.6a and to 60 m h above 1a see applications information input 5v to 15v 1578 ta01 c2 0.33 f c c 100pf d1 1n5818 c1 100 m f, 10v solid tantalum c3* 10 m f to 50 m f open = on d2 1n914 l1** 15 m h v sw fb gnd v c + + r2 4.99k r1 8.66k typical applicatio n u
2 LT1578/LT1578-2.5 parameter conditions min typ max units feedback voltage 1.195 1.21 1.225 v all conditions l 1.18 1.24 v sense voltage (fixed 2.5) 2.46 2.5 2.54 v all conditions l 2.44 2.56 v sense pin resistance 5.7 9.5 13.7 k w reference voltage line regulation 4.3v v in 15v l 0.01 0.03 %/v feedback input bias current l 0.5 2 m a error amplifier voltage gain (notes 2, 10) 200 400 error amplifier transconductance (note 10) d i (v c ) = 10 m a 800 1050 1300 m mho l 400 1700 m mho v c pin to switch current transconductance 1.5 a/ v error amplifier source current v fb = 1.1v l 40 110 190 m a error amplifier sink current v fb = 1.4v l 50 130 200 m a v c pin switching threshold duty cycle = 0 0.8 v v c pin high clamp 2.1 v switch current limit v c open, v fb = 1.1v, dc 50% l 1.5 2 3.5 a slope compensation (note 8) dc = 80% 0.3 a switch on resistance (note 7) i sw = 1.5a 0.2 0.35 w l 0.45 w maximum switch duty cycle v fb = 1.1v 90 94 % l 86 94 % minimum switch duty cycle (note 9) 8% switch frequency v c set to give 50% duty cycle 180 200 220 khz l 160 240 khz switch frequency line regulation 4.3v v in 15v l 0 0.15 %/ v frequency shifting threshold on fb pin d f = 10khz l 0.4 0.74 1.0 v minimum input voltage (note 3) l 4.0 4.3 v minimum boost voltage (note 4) i sw 1.5a l 2.3 3.0 v absolute m axi m u m ratings w ww u package/order i n for m atio n w u u (note 1) input voltage .......................................................... 16v boost pin above input voltage ............................. 10v shdn pin voltage ..................................................... 7v sense pin voltage .................................................... 4v fb pin voltage (adjustable part) ............................ 3.5v fb pin current (adjustable part) ............................ 1ma sync pin voltage ..................................................... 7v operating junction temperature range LT1578c ............................................... 0 c to 125 c LT1578i ........................................... C 40 c to 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c the l denotes specifications which apply over the full operating tempera- ture range, otherwise specifications are at t j = 25 c. v in = 5v, v c = 1.5v, boost = v in + 5v, switch open, unless otherwise noted. electrical characteristics consult factory for military grade parts. order part number LT1578cs8 LT1578is8 LT1578cs8-2.5 LT1578is8-2.5 s8 part marking 1578 1578i 1 2 3 4 8 7 6 5 top view s8 package 8-lead plastic so v sw v in boost gnd shdn fb/sense v c sync q ja =80 c/ w with fused ground pin connected to ground plane or large lands 157825 578i25
3 LT1578/LT1578-2.5 parameter conditions min typ max units boost current (note 5) i sw = 0.5a l 918 ma i sw = 1.5a l 27 50 ma v in supply current (note 6) l 1.9 2.7 ma shutdown supply current v shdn = 0v, v in 15v, v sw = 0v, v c open 20 50 m a l 75 m a lockout threshold v c open l 2.34 2.42 2.50 v shutdown thresholds v c open device shutting down l 0.13 0.37 0.60 v device starting up l 0.25 0.45 0.7 v synchronization threshold 1.5 2.2 v synchronizing range 250 400 khz sync pin input resistance 40 k w note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: gain is measured with a v c swing equal to 200mv above the switching threshold level to 200mv below the upper clamp level. note 3: minimum input voltage is not measured directly, but is guaranteed by other tests. it is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator frequency remain constant. actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. see applications information. note 4: this is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. note 5: boost current is the current flowing into the boost pin with the pin held 5v above input voltage. it flows only during switch on time. note 6: input supply current is the bias current drawn by the input pin with switching disabled. note 7: switch on resistance is calculated by dividing v in to v sw voltage by the forced current (1.5a). see typical performance characteristics for the graph of switch voltage at other currents. note 8: slope compensation is the current subtracted from the switch current limit at 80% duty cycle. see maximum output load current in the applications information section for further details. note 9: minimum on-time is 400ns typical. for a 200khz operating frequency this means the minimum duty cycle is 8%. in frequency foldback mode, the effective duty cycle will be less than 8%. note 10: transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. to calculate gain and transconductance referred to the sense pin on the fixed voltage parts, divide values shown by the ratio 2.5/1.21. typical perfor m a n ce characteristics uw switch voltage drop junction temperature ( c) ?0 1.23 1.22 1.21 1.20 1.19 100 1576 g03 25 0 25 50 75 125 feedback voltage (v) switch current (a) 0 switch voltage (v) 0.5 0.4 0.3 0.2 0.1 0 0.25 0.50 0.75 1.00 1576 g01 1.25 1.50 125 c ?0 c 25 c feedback pin voltage switch peak current limit duty cycle (%) 0 switch peak current (a) 2.5 2.0 1.5 1.0 0.5 0 80 1576 g02 20 40 60 100 typical minimum the l denotes specifications which apply over the full operating tempera- ture range, otherwise specifications are at t j = 25 c. v in = 5v, v c = 1.5v, boost = v in + 5v, switch open, unless otherwise noted. electrical characteristics
4 LT1578/LT1578-2.5 typical perfor m a n ce characteristics u w junction temperature ( c) ?0 4 3 2 1 0 100 1576 g04 25 0 25 50 75 125 shdn pin current ( a) at 2.44v standby threshold (current flows out of pin) shutdown pin bias current (v shdn = lockout threshold) junction temperature ( c) ?0 180 160 140 120 100 80 60 40 20 0 100 1576 g05 25 0 25 50 75 125 shdn pin current ( a) current required to force shutdown (flows out of pin). after shutdown, current drops to a few a junction temperature ( c) ?0 shutdown pin voltage (v) 100 1576 g06 050 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 ?5 25 75 125 start-up shutdown shutdown supply current input voltage (v) 0 input supply current ( a) 25 20 15 10 5 0 1576 g08 5 10 15 v shdn = 0v frequency (hz) gain ( m mho) phase (deg) 2000 1500 1000 500 0 500 200 150 100 50 0 ?0 10 1k 10k 1m 1576 g09 100 100k gain phase error amplifier equivalent circuit r out 570k c out 2.4pf v c r load = 50 w v fb 1 10 ? ) ( error amplifier transconductance junction temperature ( c) ?0 shutdown pin voltage (v) 2.46 2.45 2.44 2.43 2.42 2.41 2.40 25 75 1576 g07 ?5 0 50 100 125 on standby feedback voltage (v) 0 switching frequency (khz) or current ( a) 2.0 1576 g12 0.5 1.0 1.5 250 200 150 100 50 0 feedback pin current switching frequency shutdown supply current junction temperature ( c) ?0 transconductance ( mho) 100 1576 g11 050 1600 1400 1200 1000 800 600 400 200 0 25 25 75 125 error amplifier transconductance frequency foldback shutdown thresholds standby thresholds shutdown voltage (v) 0 input supply current ( a) 30 25 20 15 10 5 0 1576 g010 0.1 0.2 0.3 0.4 v in = 10v shutdown pin bias current (v shdn = shutdown threshold)
5 LT1578/LT1578-2.5 typical perfor m a n ce characteristics u w kool m m is a registered trademark of magnetics, inc. metglas is a registered trademark of alliedsignal, inc. junction temperature ( c) ?0 240 220 200 180 160 100 1576 g13 25 0 25 50 75 125 frequency (khz) switching frequency input voltage (v) 6 output current (a) 0.6 0.8 1.0 1578 g15 0.4 0.2 0 912 1.2 l = 60 h 1.4 1.6 15 l = 30 h l = 15 h maximum output current at v out = 5v load current (ma) 1 input voltage (v) 4.50 4.25 4.00 3.75 3.50 10 100 1000 1576 g14 minimum input voltage to start with 3.3v output input voltage (v) 4 output current (a) 0.6 0.8 1.0 1578 g16 0.4 0.2 0 6 8 10 12 1.2 1.4 1.6 14 l = 30 h l = 15 h l = 60 h maximum output current at v out = 3.3v input voltage (v) 4 output current (a) 0.6 0.8 1.0 1578 g17 0.4 0.2 0 6 8 10 12 1.2 1.4 1.6 14 l = 30 h l = 15 h l = 60 h maximum output current at v out = 2.5v boost pin current v c pin shutdown threshold switch current (a) 0 boost pin current (ma) 30 25 20 15 10 5 0 0.25 0.50 0.75 1.00 1576 g20 1.25 1.50 junction temperature ( c) ?0 1.0 0.8 0.6 0.4 0.2 0 100 1576 g21 25 0 25 50 75 125 threshold voltage (v) feedback pin voltage (v) 0 0 switch current limit (a) 0.5 1.0 1.5 2.0 3.0 0.2 0.4 0.6 0.8 1578 g19 1.0 1.2 2.5 switch current limit foldback
6 LT1578/LT1578-2.5 pi n fu n ctio n s uuu v sw (pin 1): the switch pin is the emitter of the on-chip power npn switch. this pin is driven up to the input pin voltage during switch on time. inductor current drives the switch pin negative during switch off time. negative volt- age is clamped with the external catch diode. maximum negative switch voltage allowed is C 0.8v. v in (pin 2): this is the collector of the on-chip power npn switch. this pin powers the internal circuitry and internal regulator. at npn switch on and off, high di/dt edges occur through this pin. keep the external bypass and catch diode close to this pin. trace inductance in this path will create a voltage spike at switch off, adding to the v ce voltage across the internal npn. boost (pin 3): the boost pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar npn power switch. without this added voltage, the typical switch voltage loss would be about 1.5v. the additional boost voltage allows the switch to saturate with its voltage drop approximating that of a 0.2 w fet struc- ture. efficiency improves from 75% for conventional bipo- lar designs to > 88% for the LT1578. gnd (pin 4): the gnd pin connection needs consideration for two reasons. first, it acts as the reference for the regulated output, so load regulation will suffer if the ground end of the load is not at the same voltage as the gnd pin of the ic. this condition will occur when load current or other currents flow through metal paths be- tween the gnd pin and the load ground point. keep the ground path short between the gnd pin and the load and use a ground plane when possible. the second consider- ation is emi caused by gnd pin current spikes. internal capacitance between the v sw pin and the gnd pin creates very narrow (<10ns) current spikes in the gnd pin. if the gnd pin is connected to system ground with a long metal trace, this trace may radiate emi. keep the path between the input bypass and the gnd pin short. the gnd pin of the so-8 package is directly attached to the internal tab. this pin should be attached to a large copper area to improve thermal resistance. v c (pin 5): the v c pin is the output of the error amplifier and the input to the peak switch current comparator. it is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. this pin sits at about 1v for very light loads and 2v at maximum load. it can be driven to ground to shut off the regulator, but if driven high, current must be limited to 4ma. fb/sense (pin 6): the feedback pin is used to set output voltage using an external voltage divider that generates 1.21v at the pin with the desired output voltage. the fixed voltage (2.5v) parts have the divider included on the chip and the fb pin is used as a sense pin, connected directly to the 2.5v output. three additional functions are per- formed by the fb pin. when the pin voltage drops below 0.7v, the switch current limit and the switching frequency are reduced and the external sync function is disabled. see feedback pin function section in applications information for details. shdn (pin 7): the shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. actually, this pin has two separate thresh- olds, one at 2.42v to disable switching, and a second at 0.4v to force complete micropower shutdown. the 2.42v threshold functions as an accurate undervoltage lockout (uvlo). this can be used to prevent the regulator from operating until the input voltage has reached a predeter- mined level. sync (pin 8): the sync pin is used to synchronize the internal oscillator to an external signal. it is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. the synchronizing range is equal to initial operating frequency, up to 400khz. when not used, this pin should be grounded. see synchronizing section in applications information for details.
7 LT1578/LT1578-2.5 block diagra m w and output capacitor, then an abrupt 180 shift will occur. the current fed system will have 90 phase shift at a much lower frequency, but will not have the additional 90 shift until well beyond the lc resonant frequency. this makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. high switch efficiency is attained by using the boost pin to provide a voltage to the switch driver which is higher than the input voltage, allowing the switch to saturate. this boosted voltage is generated with an external capacitor and diode. two comparators are connected to the shut- down pin. one has a 2.42v threshold for undervoltage lockout and the second has a 0.4v threshold for complete shutdown. the LT1578 is a constant frequency, current mode buck converter. this means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. in addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. a switch cycle starts with an oscilla- tor pulse which sets the r s flip-flop to turn the switch on. when switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. this technique means that the error amplifier commands current to be delivered to the output rather than voltage. a voltage fed system will have low phase shift up to the resonant frequency of the inductor + + + + s input 2.9v bias regulator 200khz oscillator frequency shift circuit v sw fb v c lockout comparator gnd 1578 bd slope comp 0.025 w internal v cc current sense amplifier dc voltage gain = 35 sync shdn shutdown comparator current comparator error amplifier g m = 1000 m mho foldback current limit clamp boost r s flip-flop driver circuitry s r 0.8v q2 q1 power switch 1.21v 2.42v + 0.4v 3.5 m a figure 1. block diagram
8 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u figure 2. frequency and current limit foldback + 1.21v v sw v c gnd to sync circuit 1578 f02 to frequency shifting r3 1k r4 1k r1 r2 5k output 5v r5 5k error amplifier fb 1.4v q1 LT1578 q2 + feedback pin functions the feedback (fb) pin on the LT1578 is used to set output voltage and provide several overload protection features. the first part of this section deals with selecting resistors to set output voltage and the remaining part talks about foldback frequency and current limiting created by the fb pin. please read both parts before committing to a final design. the fixed 2.5v LT1578-2.5 has internal divider resistors and the fb pin, renamed sense, is connected directly to the 2.5v output. the suggested value for the output divider resistor (see figure 2) from fb to ground (r2) is 5k or less, and a formula for r1 is shown below. the output voltage error caused by ignoring the input bias current on the fb pin is less than 0.25% with r2 = 5k. please read the following if divider resistors are increased above the suggested values. r rv out 1 2121 121 = - () . . more than just voltage feedback the feedback pin is used for more than just output voltage sensing. it also reduces switching frequency and current limit when output voltage is very low (see the frequency foldback graph in typical performance characteristics). this is done to control power dissipation in both the ic and the external diode and inductor during short-circuit con- ditions. a shorted output requires the switching regulator to operate at very low duty cycles, and the average current through the diode and inductor is equal to the short-circuit current limit of the switch (typically 2a for the LT1578, folding back to less than 0.77a). minimum switch on time limitations would prevent the switcher from attaining a sufficiently low duty cycle if switching frequency were maintained at 200khz, so frequency is reduced by about 5:1 when the feedback pin voltage drops below 0.7v (see frequency foldback graph). this does not affect operation with normal load conditions; one simply sees a gear shift in switching frequency during start-up as the output voltage rises.
9 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u in addition to lower switching frequency, the LT1578 also operates at lower switch current limit when the feedback pin voltage drops below 0.7v. q2 in figure 2 performs this function by clamping the v c pin to a voltage less than its normal 2.1v upper clamp level. this foldback current limit greatly reduces power dissipation in the ic, diode and inductor during short-circuit conditions. external synchro- nization is also disabled to prevent interference with foldback operation. again, it is nearly transparent to the user under normal load conditions. the only loads that may be affected are current sources, such as lamps and mo- tors, that maintain high load current with output voltage less than 50% of final value. in these rare situations the feedback pin can be clamped above 0.7v to defeat foldback current limit. caution: clamping the feedback pin means that frequency shifting will also be defeated, so a combina- tion of high input voltage and dead shorted output may cause the LT1578 to lose control of current limit. the internal circuitry which forces reduced switching frequency also causes current to flow out of the feedback pin when output voltage is low. the equivalent circuitry is shown in figure 2. q1 is completely off during normal operation. if the fb pin falls below 0.7v, q1 begins to conduct current and reduces frequency at the rate of approximately 1khz/ m a. to ensure adequate frequency foldback (under worst-case short-circuit conditions), the external divider thevinin resistance must be low enough to pull 35 m a out of the fb pin with 0.5v on the pin (r div 14.3k). the net result is that reductions in frequency and current limit are affected by output voltage divider imped- ance. although divider impedance is not critical, caution should be used if resistors are increased beyond the suggested values and short-circuit conditions will occur with high input voltage . high frequency pickup will increase and the protection accorded by frequency and current foldback will decrease. maximum output load current maximum load current for a buck converter is limited by the maximum switch current rating (i p ) of the LT1578. this current rating is 1.5a up to 50% duty cycle (dc), decreasing to 1.3a at 80% duty cycle. this is shown graphically in typical performance characteristics and as shown in the formula below: i p = 1.5a for dc 50% i p = 1.67 C 0.18 (dc) C 0.32(dc) 2 for 50% < dc < 90% dc = duty cycle = v out /v in example: with v out = 5v, v in = 8v; dc = 5/8 = 0.625, and; i sw(max) = 1.67 C 0.18 (0.625) C 0.32(0.625) 2 = 1.43a current rating decreases with duty cycle because the LT1578 has internal slope compensation to prevent cur- rent mode subharmonic switching. for more details, read application note 19. the LT1578 is a little unusual in this regard because it has nonlinear slope compensation which gives better compensation with less reduction in current limit. maximum load current would be equal to maximum switch current for an infinitely large inductor , but with finite inductor size, maximum load current is reduced by one-half peak-to-peak inductor current. the following formula assumes continuous mode operation, implying that the term on the right is less than one-half of i p . i out(max) = continuous mode for the conditions above and l = 15 m h, i a out max () - =- () - () ? ? ? ? ? ? () =-= 143 58 5 2 15 10 200 10 8 143 031 112 63 . ... at v in = 15v, duty cycle is 33%, so i p is just equal to a fixed 1.5a, and i out(max) is equal to: 15 515 5 2 15 10 200 10 15 15 056 094 63 . .. . - () - () ? ? ? ? ? ? () =- = - a i p - () - () ()()( ) vvv lfv out in out in 2
10 LT1578/LT1578-2.5 note that there is less load current available at the higher input voltage because inductor ripple current increases. this is not always the case. certain combinations of inductor value and input voltage range may yield lower available load current at the lowest input voltage due to reduced peak switch current at high duty cycles. if load current is close to the maximum available, please check maximum available current at both input voltage extremes. to calculate actual peak switch current with a given set of conditions, use: ii vvv lfv sw peak out out in out in ( ) =+ - () ()()( ) 2 for lighter loads where discontinuous operation can be used, maximum load current is equal to: i out(max) = discontinuous mode example: with l = 5 m h, v out = 5v, and v in(max ) = 15v, ia out max () - = () ? ? ? ? ? ? () () - () = 1 5 200 10 5 10 15 2 5 15 5 034 2 36 . . the main reason for using such a tiny inductor is that it is physically very small, but keep in mind that peak-to-peak inductor current will be very high. this will increase output ripple voltage. if the output capacitor has to be made larger to reduce ripple voltage, the overall circuit could actually wind up larger. choosing the inductor and output capacitor for most applications the output inductor will fall in the range of 15 m h to 60 m h. lower values are chosen to reduce applicatio n s i n for m atio n wu u u physical size of the inductor. higher values allow more output current because they reduce peak current seen by the LT1578 switch, which has a 1.5a limit. higher values also reduce output ripple voltage, and reduce core loss. graphs in the typical performance characteristics section show maximum output load current versus inductor size and input voltage. when choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, emi, fault cur- rent in the inductor, saturation, and of course, cost. the following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. choose a value in microhenries from the graphs of maximum load current and core loss. choosing a small inductor may result in discontinuous mode operation at lighter loads, but the LT1578 is designed to work well in either mode. keep in mind that lower core loss means higher cost, at least for closed core geometries like toroids. assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. if maxi- mum load current is 0.5a, for instance, a 0.5a inductor may not survive a continuous 1.5a overload condition. dead shorts will actually be more gentle on the induc- tor because the LT1578 has foldback current limiting. 2. calculate peak inductor current at full load current to ensure that the inductor will not saturate. peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so dont omit this step. powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. other core materials fall somewhere in between. the following formula assumes continu- ous mode of operation, but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. iflv vvv pin out in out ()()()( ) () - () 2 2
11 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u ii vvv flv peak out out in out in =+ - () ()( )( ) 2 v in = maximum input voltage f = switching frequency, 200khz 3. decide if the design can tolerate an open core geom- etry like a rod or barrel, with high magnetic field radiation, or whether it needs a closed core like a toroid to prevent emi problems. one would not want an open core next to a magnetic storage media, for instance! this is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radia- tion will be a problem. 4. start shopping for an inductor (see representative surface mount units in table 1) which meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating), and fault current (if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). keep in mind that all good things like high efficiency, low profile, and high temperature operation will increase cost, sometimes dramatically. get a quote on the cheapest unit first to calibrate yourself on price, then ask for what you really want. 5. after making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. use the experts in the linear technologys applica- tions department if you feel uncertain about the final choice. they have experience with a wide range of inductor types and can tell you about the latest devel- opments in low profile, surface mounting, etc. table 1 series core vendor/ value dc core resis- mater- height part no. ( m h) (amps) type tance( w ) ial (mm) coiltronics ctx15-2 15 1.7 tor 0.059 km m 6.0 ctx33-2 33 1.4 tor 0.106 km m 6.0 ctx68-4 68 1.2 tor 0.158 km m 6.4 ctx15-1p 15 1.4 tor 0.087 52 4.2 ctx33-2p 33 1.3 tor 0.126 52 6.0 ctx68-4p 68 1.1 tor 0.238 52 6.4 sumida cdrh74-150 15 1.47 sc 0.081 fer 4.5 cdh115-330 33 1.68 sc 0.082 fer 5.2 cdrh125-680 68 1.5 sc 0.12 fer 6 cdh74-330 33 1.45 sc 0.17 fer 5.2 coilcraft do3308p-153 15 2 sc 0.12 fer 3 do3316p-333 33 2 sc 0.1 fer 5.21 do3316p-683 68 1.4 sc 0.18 fer 5.21 pulse pe-53602 35 1.4 tor 0.166 fer 9.1 pe-53604 73 1.3 tor 0.290 fer 9.1 pe-53632 22 2.7 tor 0.063 fer 9.1 pe-53633 40 2.7 tor 0.085 fer 10 gowanda smp3316-152k 15 3.5 sc 0.041 fer 6 smp3316-332k 33 2.3 sc 0.092 fer 6 smp3316-682k 68 1.7 sc 0.178 fer 6 tor = toroid sc = semi-closed geometry fer = ferrite core material 52 = type 52 powdered iron core material km m = kool m m
12 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u output capacitor ripple current (rms): i vvv lfv ripple rms out in out in ( ) = () - () ()()( ) 029 . ceramic capacitors higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. these are tempt- ing for switching regulator use because of their very low esr. unfortunately, the esr is so low that it can cause loop stability problems. solid tantalum capacitors esr generates a loop zero at 5khz to 50khz that is instrumen- tal in giving acceptable loop phase margin. ceramic capacitors remain capacitive to beyond 300khz and usu- ally resonate with their esl before their esr provides any damping. they are appropriate for input bypassing be- cause of their high ripple current ratings and tolerance of turn-on surges. output ripple voltage figure 3 shows a typical output ripple voltage waveform for the LT1578. ripple voltage is determined by the high frequency impedance of the output capacitor, and ripple current through the inductor. peak-to-peak ripple current through the inductor into the output capacitor is: i vvv vlf p out in out in -p = () - () ()()() for high frequency switchers, the sum of ripple current slew rates may also be relevant and can be calculated from: s di dt v l in = output capacitor the output capacitor is normally chosen by its effective series resistance (esr), because this is what determines output ripple voltage. to get low esr takes volume , so physically smaller capacitors have high esr. the esr range for typical LT1578 applications is 0.05 w to 0.2 w . a typical output capacitor is an avx type tps, 100 m f at 10v, with a guaranteed esr less than 0.1 w . this is a d size surface mount solid tantalum capacitor. tps capacitors are specially constructed and tested for low esr, so they give the lowest esr for a given volume. the value in microfarads is not particularly critical, and values from 22 m f to greater than 500 m f work well, but you cannot cheat mother nature on esr. if you find a tiny 22 m f solid tantalum capacitor, it will have high esr, and output ripple voltage will be terrible. table 2 shows some typical solid tantalum surface mount capacitors. table 2. surface mount solid tantalum capacitor esr and ripple current e case size esr (max., w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.7 to 0.9 0.4 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 c case size avx tps 0.2 (typ) 0.5 (typ) many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. this is historically true, and type tps capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. high discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. unlike the input capacitor, rms ripple current in the output capacitor is normally low enough that ripple cur- rent rating is not an issue. the current waveform is triangular with a typical value of 200ma rms . the formula to calculate this is:
13 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u catch diode the suggested catch diode (d1) is a 1n5818 schottky, or its motorola equivalent, mbr130. it is rated at 1a average forward current and 30v reverse voltage. typical forward voltage is 0.42v at 1a. the diode conducts current only during switch off time. peak reverse voltage is equal to regulator input voltage. average forward current in normal operation can be calculated from: 2 m s/div 1578 f03 peak-to-peak output ripple voltage is the sum of a triwave created by peak-to-peak ripple current times esr, and a square wave created by parasitic inductance (esl) and ripple current slew rate. capacitive reactance is assumed to be small compared to esr or esl. v i esr esl di dt ripple = ()( ) + () p-p s example: with v in =10v, v out = 5v, l = 30 m h, esr = 0.1 w , esl = 10nh: ia di dt va mv ripple p-p p-p = () - () () ? ? ? ? ? ? = == = ()() + ? ? ? ? ? ? =+= - - - 510 5 10 30 10 200 10 042 10 30 10 033 10 0 42 0 1 10 10 0 33 10 0 042 0 003 45 63 6 6 96 . . .. . .. s i ivv v d avg out in out in ( ) = - () this formula will not yield values higher than 1a with maximum load current of 1.25a unless the ratio of input to output voltage exceeds 5:1. the only reason to consider a larger diode is the worst-case condition of a high input voltage and overloaded (not shorted) output. under short- circuit conditions, foldback current limit will reduce diode current to less than 1a, but if the output is overloaded and does not fall to less than 1/3 of nominal output voltage, foldback will not take effect. with the overloaded condi- tion, output current will increase to a typical value of 1.8a, determined by peak switch current limit of 2a. with v in = 15v, v out = 4v (5v overloaded) and i out = 1.8a: ia d avg () = - () = 1 8 15 4 15 132 . . this is safe for short periods of time, but it would be prudent to check with the diode manufacturer if continu- ous operation under these conditions must be tolerated. boost pin considerations for most applications, the boost components are a 0.33 m f capacitor and a 1n914 or 1n4148 diode. the anode is connected to the regulated output voltage and this gener- ates a voltage across the boost capacitor nearly identical to the regulated output. in certain applications, the anode may instead be connected to the unregulated input volt- age. this could be necessary if the regulated output voltage is very low (< 3v) or if the input voltage is less than 6v. efficiency is not affected by the capacitor value, but the capacitor should have an esr of less than 1 w to ensure that it can be recharged fully under the worst-case condi- tion of minimum input voltage. almost any type of film or ceramic capacitor will work fine. warning! peak voltage on the boost pin is the sum of unregulated input voltage plus the voltage across the v out at i out = 1a inductor current at i out = 1a v out at i out = 50ma inductor current at i out = 50ma 20mv/div 200ma/div 20mv/div 200ma/div figure 3. LT1578 ripple voltage waveform
14 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u boost capacitor. this normally means that peak boost pin voltage is equal to input voltage plus output voltage, but when the boost diode is connected to the regulator input, peak boost pin voltage is equal to twice the input voltage. be sure that boost pin voltage does not exceed its maximum rating . for nearly all applications, a 0.33 m f boost capacitor works just fine, but for the curious, more details are provided here. the size of the boost capacitor is determined by switch drive current requirements. during switch on time, drain current on the capacitor is approximately i out / 50. at peak load current of 1.25a, this gives a total drain of 25ma. capacitor ripple voltage is equal to the product of on time and drain current divided by capacitor value; d v = (t on )(25ma/c). to keep capacitor ripple voltage to less than 0.5v (a slightly arbitrary number) at the worst- case condition of t on = 4.7 m s, the capacitor needs to be 0.24 m f. boost capacitor ripple voltage is not a critical parameter, but if the minimum voltage across the capaci- tor drops to less than 3v, the power switch may not saturate fully and efficiency will drop. an approximate formula for absolute minimum capacitor value is: + + 2.42v 0.4v gnd v sw LT1578 input r fb r hi 1578 f04 output shdn standby in total shutdown 3.5 m a r lo c1 + figure 4. undervoltage lockout c ivv fv v min out out in out = ()( ) () - () // 50 3 f = switching frequency v out = regulated output voltage v in = minimum input voltage this formula can yield capacitor values substantially less than 0.24 m f, but it should be used with caution since it does not take into account secondary factors such as capacitor series resistance, capacitance shift with tem- perature and output overload. shutdown function and undervoltage lockout figure 4 shows how to add undervoltage lockout (uvlo) to the LT1578. typically, uvlo is used in situations where the input supply is current limited , or has a relatively high source resistance. it is particularly useful for input sup- plies with foldback current limiting. a switching regulator draws constant power from the source, so source current increases as source voltage drops. this looks like a negative resistance load to the source and can cause the source to current limit and latch under low source voltage
15 LT1578/LT1578-2.5 conditions. uvlo helps prevent the regulator from oper- ating at source voltages where these problems might occur. threshold voltage for lockout is about 2.42v. a 3.5 m a bias current flows out of the pin at threshold. this internally generated current is used to force a default high state on the shutdown pin if the pin is left open. when low shut- down current is not an issue, the error due to this current can be minimized by making r lo 10k or less. if shutdown current is an issue, r lo can be raised to 100k, but the error due to initial bias current and changes with temperature should be considered. rk r rv v vr a lo hi lo in lo = () = - () - () 10 242 242 35 to 100k 25k suggested . .. m v in = minimum input voltage keep the connections from the resistors to the shutdown pin short and make sure that interplane or surface capaci- tance to the switching nodes are minimized. if high resis- tor values are used, the shutdown pin should be bypassed with a 1000pf capacitor to prevent coupling problems from the switch node. if hysteresis is desired in the undervoltage lockout point, a resistor r fb can be added to the output node. resistor values can be calculated from: r rv vv v r a rrv v hi lo in out lo fb hi out = -+ () + [] - () = ()( ) 2 42 1 242 35 ./ .. / dd d m 25k suggested for r lo v in = input voltage at which switching stops as input voltage descends to trip level d v = hysteresis in input voltage level example: output voltage is 5v, switching is to stop if input voltage drops below 12v and should not restart unless applicatio n s i n for m atio n wu u u input rises back to 13.5v. d v is therefore 1.5v and v in = 12v. let r lo = 25k. r k ka k k rk k hi fb = -+ () + [] - () = () = = () = 25 12 2 42 15 5 1 15 242 25 35 25 10 35 233 111 111 5 1 5 370 ../ . .. . . /. m switch node considerations for maximum efficiency, switch rise and fall times are made as short as possible. to prevent radiated emi and high frequency resonance problems, proper layout of the components connected to the switch node is essential. b field (magnetic) radiation is minimized by keeping catch diode, switch pin, and input bypass capacitor leads as short as possible. e field radiation is kept low by minimiz- ing the length and area of all traces connected to the switch pin and boost pin. a ground plane should always be used under the switcher circuitry to prevent interplane cou- pling. a suggested layout for the critical components is shown in figure 5. note that the feedback resistors and compensation components are kept as far as possible from the switch node. also note that the high current ground path of the catch diode and input capacitor are kept very short and separate from the analog ground line. the high speed switching current path is shown schemati- cally in figure 6. minimum lead length in this path is essential to ensure clean switching and low emi. the path including the switch, catch diode, and input capacitor is the only one containing nanosecond rise and fall times. if you follow this path on the pc layout, you will see that it is irreducibly short. if you move the diode or input capacitor away from the LT1578, get your resum in order. the other paths contain only some combination of dc and 200khz triwave, so are much less critical.
16 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u figure 6. high speed switching path figure 5. suggested layout for LT1578 v out v in sw boost fb sync shdn v c gnd 1578 f05 gnd keep input capacitor and catch diode close to regulator and terminate them to the same point connect output capacitor directly to heavy ground take output directly from end of output capacitor to avoid parasitic resistance and inductance (kelvin connection) minimize area of connections to switch node and boost node ground ring need not be as shown (normally exists as internal plane) minimize size of feedback pin connections to avoid pickup terminate feedback resistors and compensation components directly to switcher ground pin c c r c r1 d1 c3 d2 l1 c1 c2 r2 1578 f06 5v l1 v in high frequency circulating path load switch node
17 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u parasitic resonance resonance or ringing may sometimes be seen on the switch node (see figure 7). very high frequency ringing following the switch voltage rise time is caused by switch/ diode/input capacitance lead inductance and diode ca- pacitance. schottky diodes have very high q junction capacitance that can ring for many cycles when excited at high frequency. if total lead length for the input capacitor, diode and switch path is 1 inch, the inductance will be approximately 25nh. at switch off, this will produce a spike across the npn output device in addition to the input voltage. at higher currents this spike can be in the order of 10v to 20v or higher with a poor layout, potentially exceeding the absolute max switch voltage. the path around switch, catch diode and input capacitor must be kept as short as possible to ensure reliable operation. when looking at this, a >100mhz oscilloscope must be used, and waveforms should be observed on the leads of the package. this switch off spike will also cause the sw node to go below ground. the LT1578 has special circuitry inside which mitigates this problem, but negative voltages over 1v lasting longer than 10ns should be avoided. note that 100mhz oscilloscopes are barely fast enough to see the details of the falling edge overshoot in figure 7. a second, much lower frequency ringing is seen during switch off time if load current is low enough to allow the inductor current to fall to zero during part of the switch off time (see figure 8). switch and diode capacitance reso- nate with the inductor to form damped ringing at 1mhz to 10 mhz. this ringing is not harmful to the regulator and it has not been shown to contribute significantly to emi. any attempt to damp it with an rc snubber will slightly degrade efficiency. input bypassing and voltage range input bypass capacitor step-down converters draw current from the input supply in pulses. the average height of these pulses is equal to load current, and the duty cycle is equal to v out /v in . rise and fall times of the current are very fast. a local bypass capacitor across the input supply is necessary to ensure proper operation of the regulator and minimize the ripple current fed back into the input supply. the capacitor also forces switching current to flow in a tight local loop, minimizing emi . do not cheat on the ripple current rating of the input bypass capacitor, but also do not be overly concerned with the value in microfarads . the input capacitor is intended to absorb all the switching current ripple, which can have an rms value as high as one half of the load current. ripple current ratings on the capacitor must be observed to ensure reliable operation. in many cases it is necessary to parallel two capacitors to obtain the required ripple rating. both capacitors must be of the same value and manufac- turer to guarantee power sharing. the actual value of the capacitor in microfarads is not particularly important figure 7. switch node response figure 8. discontinuous mode ringing 5v/div 50ma/div 50ns/div 1578 f07 1 m s/div 1578 f08 inductor current switch node voltage rise and fall waveforms are superimposed (pulse width is not 350ns) 5v/div
18 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u because at 200khz, any value above 15 m f is essentially resistive. rms ripple current rating is the critical param- eter. actual rms current can be calculated from: iivvvv ripple rms out out in out in () =- () / 2 the term inside the radical has a maximum value of 0.5 when input voltage is twice output, and stays near 0.5 for a relatively wide range of input voltages. it is common practice therefore to simply use the worst-case value and assume that rms ripple current is one half of load current. at maximum output current of 1.5a for the LT1578, the input bypass capacitor should be rated at 0.75a ripple current. note however, that there are many secondary considerations in choosing the final ripple current rating. these include ambient temperature, average versus peak load current, equipment operating schedule, and required product lifetime. for more details, see application notes 19 and 46, and design note 95. input capacitor type some caution must be used when selecting the type of capacitor used at the input to regulators. aluminum electrolytics are lowest cost, but are physically large to achieve adequate ripple current rating, and size con- straints (especially height) may preclude their use. ceramic capacitors are now available in larger values, and their high ripple current and voltage rating make them ideal for input bypassing. cost is fairly high and footprint may also be somewhat large. solid tantalum capacitors would be a good choice, except that they have a history of occasional spectacular failures when they are subjected to large current surges during power-up. the capacitors can short and then burn with a brilliant white light and lots of nasty smoke. this phenomenon occurs in only a small percentage of units, but it has led some oems to forbid their use in high surge applications. the input bypass capacitors of regulators can see these high surges when a battery or high capacitance source is connected. several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (avx tps series for instance, see table 3), but even these units may fail if the input voltage surge approaches the maximum voltage rating of the capacitor. avx recommends derating capacitor voltage by 2:1 for high surge applications. the highest voltage rating is 50v, so 25v may be a practical input voltage upper limit when using solid tantalum ca- pacitors for input bypassing. larger capacitors may be necessary when the input volt- age is very close to the minimum specified on the data sheet. small voltage dips during switch on time are not normally a problem, but at very low input voltage they may cause erratic operation because the input voltage drops below the minimum specification. problems can also occur if the input-to-output voltage differential is near minimum. the amplitude of these dips is normally a function of capacitor esr and esl because the capacitive reactance is small compared to these terms. esr tends to be the dominate term and is inversely related to physical capacitor size within a given capacitor type. synchronizing the sync pin is used to synchronize the internal oscillator to an external signal. the sync input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 10% and 90%. the input can be driven directly from a logic level output. the synchronizing range is equal to initial operating frequency up to 400khz. this means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (250khz), not the typical operating frequency of 200khz. caution should be used when synchronizing above 280khz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. this type of subharmonic switching only occurs at input voltages less than twice output voltage. higher inductor values will tend to eliminate this problem. see frequency compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. application note 19 has more details on the theory of slope compensation.
19 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u at power-up, when v c is being clamped by the fb pin (see figure 2, q2), the sync function is disabled. this allows the frequency foldback to operate in the shorted output con- dition. during normal operation, switching frequency is controlled by the internal oscillator until the fb pin reaches 0.7v, after which the sync pin becomes operational. if no synchronization is required, this pin should be connected to ground. thermal calculations power dissipation in the LT1578 chip comes from four sources: switch dc loss, switch ac loss, boost circuit current, and input quiescent current. the following formu- las show how to calculate each of these losses. these formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. switch loss: p ri v v ns i v f sw sw out out in out in = ()( ) + ()()() 2 60 boost current loss: p vi v boost out out in = () 2 50 / quiescent current loss: pv v v v q in out out in = ? ? ? + ? ? ? + ? ? ? ? () -- 055 10 16 10 0 004 33 2 . . . r sw = switch resistance ( ? 0.2 w ) 60ns = equivalent switch current/voltage overlap time f = switch frequency example: with v in = 10v, v out = 5v and i out = 1a: p w pw p sw boost q = ()()() + ? ? ? ()( ) ? ? ? =+ = = ()( ) = = ? ? ? + ? ? ? + ()( ) = - -- 02 1 5 10 60 10 1 10 2 00 10 01 012 0 22 5150 10 005 10 0 55 10 5 1 6 10 5 0 004 10 0 2 93 2 33 2 . .. . / . . . . .. 02w total power dissipation is 0.22 + 0.05 + 0.02 = 0.29w. thermal resistance for LT1578 package is influenced by the presence of internal or backside planes. with a full plane under the so package, thermal resistance will be about 80 c/w. no plane will increase resistance to about 120 c/w. to calculate die temperature, add in worst-case ambient temperature: t j = t a + q ja (p tot ) with the so-8 package ( q ja = 80 c/w), at an ambient temperature of 50 c, t j = 50 + 80 (0.29) = 73.2 c die temperature is highest at low input voltage, so use lowest continuous input operating voltage for thermal calculations. frequency compensation loop frequency compensation of switching regulators can be a rather complicated problem because the reactive components used to achieve high efficiency also intro- duce multiple poles into the feedback loop. the inductor and output capacitor on a conventional step-down con- verter actually form a resonant tank circuit that can exhibit peaking and a rapid 180 phase shift at the resonant frequency. by contrast, the LT1578 uses a current mode architecture to help alleviate the phase shift created by the inductor. the basic connections are shown in figure 9. figure 10 shows a bode plot of the phase and gain of the power section of the LT1578, measured from the v c pin to
20 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u the output. gain is set by the 1.5a/v transconductance of the LT1578 power section and the effective complex impedance from output to ground. gain rolls off smoothly above the 160hz pole frequency set by the 100 m f output capacitor. phase drop is limited to about 85 . phase recovers and gain levels off at the zero frequency ( ? 16khz) set by capacitor esr (0.1 w ). error amplifier transconductance phase and gain are shown in figure 11. the error amplifier can be modeled as a transconductance of 1000 m mho, with an output imped- ance of 570k w in parallel with 2.4pf. in all practical applications, the compensation network from the v c pin to ground has a much lower impedance than the output impedance of the amplifier at frequencies above 200hz. this means that the error amplifier characteristics them- selves do not contribute excess phase shift to the loop, and the phase/gain characteristics of the error amplifier sec- tion are completely controlled by the external compensa- tion network. in figure 12, full loop phase/gain characteristics are shown with a compensation capacitor of 100pf, giving the error amplifier a pole at 2.8khz, with phase rolling off to 90 and staying there. the overall loop has a gain of 66db at low frequency, rolling off to unity-gain at 58khz. the phase plot shows a two-pole characteristic until the esr of the output capacitor brings it back to single pole above 16khz. phase margin is about 77 at unity-gain. frequency (hz) gain ( m mho) phase (deg) 2000 1500 1000 500 0 500 200 150 100 50 0 ?0 10 1k 10k 1m 1578 f11 100 100k gain phase r out 570k c out 2.4pf v c error amplifier equivalent circuit r load = 50 w v fb 1 10 ? ) ( + 1.21v v sw v c LT1578 gnd 1578 f09 r1 output esr c f c c r c error amplifier fb r2 c1 current mode power stage g m = 1.5a/v + figure 10. response from v c pin to output frequency (hz) 10 gain (db) phase (deg) 40 20 0 ?0 ?0 40 0 ?0 ?0 120 100 1k 1578 f07 10k 100k gain phase v in = 10v v out = 5v i out = 500ma figure 12. overall loop characteristics frequency (hz) loop gain (db) loop phase (deg) 80 60 40 20 0 ?0 180 135 90 45 0 ?5 10 1k 10k 1m 1578 f12 100 100k v in = 10v v out = 5v i out = 500ma c out = 100 f 10v, avx tps c c = 100pf l = 30 h phase gain figure 9. model for loop response figure 11. error amplifier gain and phase
21 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u analog experts will note that around 7khz, phase dips close to the zero phase margin line. this is typical of switching regulators, especially those that operate over a wide range of loads. this region of low phase is not a problem as long as it does not occur near unity-gain. in practice, the variability of output capacitor esr tends to dominate all other effects with respect to loop response. variations in esr will cause unity-gain to move around, but at the same time phase moves with it so that adequate phase margin is maintained over a very wide range of esr ( 3 3:1). what about a resistor in the compensation network? it is common practice in switching regulator design to add a zero to the error amplifier compensation to increase loop phase margin. this zero is created in the external network in the form of a resistor (r c ) in series with the compensation capacitor. increasing the size of this resis- tor generally creates better and better loop stability, but there are two limitations on its value. first, the combina- tion of output capacitor esr and a large value for r c may cause loop gain to stop rolling off altogether, creating a gain margin problem. an approximate formula for r c where gain margin falls to zero is: r loop v g g esr c out mp ma gain = 1 () = ()()()() 121 . g mp = transconductance of power stage = 1.5a/v g ma = error amplifier transconductance = 1(10 C3 ) esr = output capacitor esr 1.21 = reference voltage with v out = 5v and esr = 0.1 w , a value of 27.5k for r c would yield zero gain margin, so this represents an upper limit. there is a second limitation however which has nothing to do with theoretical small signal dynamics. this resistor sets high frequency gain of the error amplifier, including the gain at the switching frequency. if the switching frequency gain is high enough, an excessive amout of output ripple voltage will appear at the v c pin resulting in improper operation of the regulator. in a marginal case, subharmonic switching occurs, as evidenced by alternating pulse widths seen at the switch node. in more severe cases, the regulator squeals or hisses audibly even though the output voltage is still roughly correct. none of this will show on a bode plot since this is an amplitude insensitive measurement. tests have shown that if ripple voltage on the v c is held to less than 100mv p-p , the LT1578 will generally be well behaved . the formula below will give an estimate of v c ripple voltage when r c is added to the loop, assuming that r c is large compared to the reactance of c c at 200khz. v r g v v esr vlf c ripple c ma in out in () = ()( ) - ()()() ()()() 121 . g ma = error amplifier transconductance (1000 m mho) if a series compensation resistor of 15k gave the best overall loop response, with adequate gain margin, the resulting v c pin ripple voltage with v in = 10v, v out = 5v, esr = 0.1 w , l = 30 m h, would be: v k v c ripple () - - = () () - ()()() () ()() = 15 1 10 10 5 01 121 10 30 10 200 10 0 151 3 63 .. . this ripple voltage is high enough to possibly create subharmonic switching. in most situations a compromise value (< 10k in this case) for the resistor gives acceptable phase margin and no subharmonic problems. in other cases, the resistor may have to be larger to get acceptable phase response, and some means must be used to control ripple voltage at the v c pin. the suggested way to do this is to add a capacitor (c f ) in parallel with the r c /c c network on the v c pin. the pole frequency for this capacitor is typically set at one-fifth of the switching frequency so that it provides significant attenuation of the switching ripple, but does not add unacceptable phase shift at the loop unity-gain frequency. with r c = 15k, c fr k pf f c = ()()() = () () = 5 2 5 2 200 10 15 265 3 p p
22 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u one way to check switching regulator loop stability is by pulse loading the regulator output while observing the transient response at the output, using the circuit shown in figure 13. the regulator loop is hit with a small transient ac load current at a relatively low frequency, 50hz to 1khz. this causes the output to jump a few millivolts, then settle back to the original value, as shown in figure 14. a well behaved loop will settle back cleanly, whereas a loop with poor phase or gain margin will ring as it settles. the number of rings indicates the degree of stability, and the frequency of the ringing shows the approximate unity-gain frequency of the loop. amplitude of the signal is not particularly important, as long as the amplitude is not so high that the loop behaves nonlinearly. how do i test loop stability? the standard compensation for LT1578 is a 100pf capacitor for c c , with r c = 0. while this compensation will work for most applications, the optimum value for loop compensation components depends, to various extents, on parameters which are not well controlled. these in- clude inductor value ( 30% due to production tolerance, load current and ripple current variations), output capaci- tance ( 20% to 50% due to production tolerance, temperature, aging and changes at the load), output capacitor esr ( 200% due to production tolerance, temperature and aging), and finally, dc input voltage and output load current . this makes it important for the designer to check out the final design to ensure that it is robust and tolerant of all these variations. 0.2ms/div 1578 f14 10mv/div v out at i out = 500ma before filter v out at i out = 500ma after filter load pulse through 50 w f ? 780hz 5a/div v out at i out = 50ma after filter figure 14. loop stability check to oscilloscope sync adjustable dc load adjustable input supply 100hz to 1khz 100mv to 1v p-p 100 m f to 1000 m f ripple filter 1578 f13 to x1 oscilloscope probe 3300pf 330pf 50 w 470 w 4.7k switching regulator + figure 13. loop stability test circuit
23 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u the output of the regulator contains both the desired low frequency transient information and a reasonable amount of high frequency (200khz) ripple. the ripple makes it difficult to observe the small transient, so a two-pole, 100khz filter has been added. this filter is not particularly critical; even if it attenuated the transient signal slightly, this wouldnt matter because amplitude is not critical. after verifying that the setup is working correctly, start varying load current and input voltage to see if you can find any combination that makes the transient response look suspiciously ringy. this procedure may lead to an ad- justment for best loop stability or faster loop transient response. nearly always you will find that loop response looks better if you add in several k w for r c . do this only if necessary, because as explained before, r c above 1k may require the addition of c f to control v c pin ripple. if everything looks ok, use a heat gun and cold spray on the circuit (especially the output capacitor) to bring out any temperature-dependent characteristics. keep in mind that this procedure does not take initial component tolerance into account. you should see fairly clean response under all load and line conditions to ensure that component variations will not cause problems. one note here: according to murphy, the component most likely to be changed in production is the output capacitor, because that is the component most likely to have manu- facturer variations (in esr) large enough to cause prob- lems. it would be a wise move to lock down the sources of the output capacitor in production. also, try varying com- ponent values by a factor of 2 and see if the behavior is still acceptable. double and halve the values of r c and c c and output capacitors. if the regulator still works correctly, it will likely be good in production. a possible exception to the clean response rule is at very light loads, as evidenced in figure 14 with i load = 50ma. switching regulators tend to have dramatic shifts in loop response at very light loads, mostly because the inductor current becomes discontinuous. one common result is very slow but stable characteristics. a second possibility is low phase margin, as evidenced by ringing at the output with transients. the good news is that the low phase margin at light loads is not particularly sensitive to component varia- tion, so if it looks reasonable under a transient test, it will probably not be a problem in production. note that fre- quency of the light load ringing may vary with component tolerance but phase margin generally hangs in there. positive-to-negative converter the circuit in figure 15 is a classic positive-to-negative topology using a grounded inductor. it differs from the standard approach in the way the ic chip derives its feedback signal. because the LT1578 accepts only posi- tive feedback signals, the ground pin must be tied to the regulated negative output. a resistor divider to ground or, in this case, the sense pin, then provides the proper feedback voltage for the chip. figure 15. positive-to-negative converter output** 5v, 0.5a input 5.5v to 15v 1578 f15 c2 0.33 f c c r c d2 1n5818 c1 100 f 10v tant 2 r1 15.8k r2 4.99k c3 10 f to 50 f d1 1n4148 l1* 15 h boost LT1578 v in v sw fb gnd v c * increase l1 to 30 h or 60 h for higher current applications. see applications information ** maximum load current depends on minimum input voltage and inductor size. see applications information + + inverting regulators differ from buck regulators in the basic switching network. current is delivered to the output as square waves with a peak-to-peak amplitude much greater than load current . this means that maximum load current will be significantly less than the LT1578s 1.5a maximum switch current, even with large inductor values . the buck converter in comparison, delivers current to the output as a triangular wave superimposed on a dc level equal to load current, and load current can approach 1.5a
24 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u with large inductors. output ripple voltage for the positive- to-negative converter will be much higher than a buck converter. ripple current in the output capacitor will also be much higher. the following equations can be used to calculate operating conditions for the positive-to-negative converter. maximum load current: i i vv vvfl vv vv vv max p in out out in out in out in out f = - ()( ) + ()()() ? ? () - () +- () + () 2 035 035 . . i p = maximum rated switch current v in = minimum input voltage v out = output voltage v f = catch diode forward voltage 0.35 = switch voltage drop at 1.5a example: with v in(min) = 5.5v, v out = 5v, l = 30 m h, v f = 0.5v, i p = 1.5a: i max = 0.6a. note that this equation does not take into account that maximum rated switch current (i p ) on the LT1578 is reduced slightly for duty cycles above 50%. if duty cycle is expected to exceed 50% (input voltage less than output voltage), use the actual i p value from the electrical characteristics table. operating duty cycle: dc vv vvv out f in out f = + -+ + 03 . (this formula uses an average value for switch loss, so it may be several percent in error.) with the conditions above: dc = + -++ = 505 55 03 5 05 51 . .. . % this duty cycle is close enough to 50% that i p can be assumed to be 1.5a. output divider if the adjustable part is used, the resistor connected to v out (r2) should be set to approximately 5k. r1 is calculated from: r rv out 1 2121 121 = - () . . inductor value unlike buck converters, positive-to-negative converters cannot use large inductor values to reduce output ripple voltage. at 200khz, values larger than 75 m h make almost no change in output ripple. the graph in figure 16 shows peak-to-peak output ripple voltage for a 5v to C 5v con- verter versus inductor value. the criteria for choosing the inductor size ( h) 0 output ripple voltage (mv p-p ) 150 120 90 60 30 0 60 1578 f16 15 30 45 75 discontinuous i load = 0.25a discontinuous i load = 0.1a 5v to 5v converter output capacitor? esr = 0.1 continuous i load > 0.38a figure 16. ripple voltage on positive-to-negative converter
25 LT1578/LT1578-2.5 applicatio n s i n for m atio n wu u u for the example above, with maximum load current of 0.25a: ia cont = ()() + () ++ () = 55 15 455555505 038 22 .. .. . . this says that discontinuous mode can be used and the minimum inductor needed is found from: lh min = ()( ) ? ? ? () = 25 025 200 10 1 5 56 3 2 . . .m in practice, the inductor should be increased by about 30% over the calculated minimum to handle losses and varia- tions in value. this suggests a minimum inductor of 7.3 m h for this application, but looking at the ripple voltage chart shows that output ripple voltage could be reduced by a fac- tor of two by using a 30 m h inductor. there is no rule of thumb here to make a final decision. if modest ripple is needed and the larger inductor does the trick, this is probably the best solution. if ripple is noncritical use the smaller inductor. if ripple is extremely critical, a second stage filter may have to be added in any case, and the lower value of inductance can be used. keep in mind that the output capacitor is the other critical factor in determining output ripple voltage. ripple shown on the graph (figure 16) is with a capacitors esr of 0.1 w . this is reasonable for avx type tps d or e size surface mount solid tantalum capacitors, but the final capacitor chosen must be looked at carefully for esr characteristics. inductor is therefore typically based on ensuring that peak switch current rating is not exceeded. this gives the lowest value of inductance that can be used, but in some cases (lower output load currents) it may give a value that creates unnecessarily high output ripple voltage. a com- promise value is often chosen that reduces output ripple. as you can see from the graph, large inductors will not give arbitrarily low ripple, but small inductors can give high ripple. the difficulty in calculating the minimum inductor size needed is that you must first know whether the switcher will be in continuous or discontinuous mode at the critical point where switch current is 1.5a. the first step is to use the following formula to calculate the load current where the switcher must use continuous mode. if your load current is less than this, use the discontinuous mode formula to calculate the minimum inductor value needed. if the load current is higher, use the continuous mode formula. output current where continuous mode is needed: i vi vv vv v cont in p in out in out f = ()() + () ++ () 22 4 minimum inductor discontinuous mode: l vi fi min out out p = ()() ()( ) 2 2 minimum inductor continuous mode: l vv fv v i i vv v min in out in out p out out f in = ()( ) () + () -+ + () ? ? ? ? ? ? ? 21
26 LT1578/LT1578-2.5 ripple current in the input and output capacitors positive-to-negative converters have high ripple current in both the input and output capacitors. for long capacitor lifetime, the rms value of this current must be less than the high frequency ripple current rating of the capacitor. the following formula will give an approximate value for rms ripple current. this formula assumes continuous conduction mode and a large inductor value . small induc- tors will give somewhat higher ripple current, especially in discontinuous mode. the exact formulas are very com- plex and appear in application note 44, pages 30 and 31. for our purposes here, a simple fudge factor (ff) is added. the value for ff is about 1.2 for load currents above 0.38a (in continuous conduction mode) and l 3 10 m h. it in- creases to about 2.0 for smaller inductors at lower load currents (in discontinuous conduction mode). capacitor ff i v v out out in i rms = ()( ) ff = fudge factor (1.2 to 2.0) applicatio n s i n for m atio n wu u u diode current average diode current is equal to load current. peak diode current will be considerably higher. peak diode current: continuous i vv v vv lfv v discontinuous v lf out in out in in out in out out mode mode = 2i out = + () + ()( ) ()() + () ()( ) ()() 2 keep in mind that during start-up and output overloads, the average diode current may be much higher than with normal loads. care should be used if diodes rated less than 1a are used, especially if continuous overload conditions must be tolerated.
27 LT1578/LT1578-2.5 dimensions in inches (millimeters) unless otherwise noted. package descriptio n u s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * ** information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
28 LT1578/LT1578-2.5 part number description comments lt1074/lt1076 step-down switching regulators 40v input, 100khz, 5a and 2a ltc1174 high efficiency step-down and inverting dc/dc converter 0.5a, 150khz burst mode tm operation lt1370 high efficiency dc/dc converter 42v, 6a, 500khz switch lt1371 high efficiency dc/dc converter 35v, 3a, 500khz switch lt1372/lt1377 500khz and 1mhz high efficiency 1.5a switching regulators boost topology lt1376 high efficiency step-down switching regulator 25v, 1.5a, 500khz switch lt1507 high efficiency step-down switching regulator 15v, 1.5a, 500khz switch lt1676/lt1776 high efficiency step-down switching regulators 7.4v to 60v input, 100khz/200khz ltc1772 sot-23 low voltage step-down dc/dc controller 550khz, drives pfet, 6-lead sot-23 package; up to 4.5a output current ltc1735 high efficiency step-down converter synchronous buck controller drives external mosfets lt1777 low noise step-down switching regulator 48v input, internally limited dv/dt, programmable di/dt burst mode is a trademark of linear technology corporation. 1578f lt/tp 0100 4k ? printed in usa ? linear technology corporation 1999 typical applicatio n u dual output sepic converter the circuit in figure 17 generates both positive and negative 5v outputs with a single piece of magnetics. the inductor l1 is a 33 m h surface mount inductor from coiltronics. it is manufactured with two identical windings that can be connected in series or parallel. the topology for the 5v output is a standard buck converter. the C 5v topology would be a simple flyback winding coupled to the buck converter if c4 were not present. c4 creates the sepic (single-ended primary inductance converter) to- pology which improves regulation and reduces ripple current in l1. without c4, the voltage swing on l1b compared to l1a would vary due to relative loading and linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com boost LT1578 v in output 5v output ?v ? * l1 is a single core with two windings coiltronics ctx33-2 ** avx tspd107m010 ? if load can go to zero, an optional preload of 1k to 5k may be used to improve load regulation input 6v to 15v gnd 1578 f17 c2 0.33 m f c c 100pf d1 1n5818 c1** 100 m f 10v tant c5** 100 m f 10v tant c3 22 m f 35v tant c4** 100 f d2 1n914 r1 15.8k r2 4.99k d3 1n5818 l1a* 33 m h l1b* v sw fb gnd shdn v c + + + + related parts figure 17. dual output sepic converter coupling losses. c4 provides a low impedance path to maintain an equal voltage swing in l1b, improving regu- lation. in a flyback converter, during switch on time, all the converters energy is stored in l1a only, since no current flows in l1b. at switch off, energy is transferred by magnetic coupling into l1b, powering the C 5v rail. c4 pulls l1b positive during switch on time, causing current to flow, and energy to build in l1b and c4. at switch off, the energy stored in both l1b and c4 supply the C 5v rail. this reduces the current in l1a and changes l1b current waveform from square to triangular. for details on this circuit see design note 100.

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